Supply circuits, such as for example, AC/DC or DC/DC switching power supplies, are well known in the art. There are many types of electronic converters that may be divided mainly into insulated and non-insulated converters. For example, non-insulated electronic converters are converters of the buck, boost, buck-boost, Cuk, SEPIC, and ZETA types. Insulated converters are, for example, converters of the flyback, forward, half-bridge, and full-bridge types. These types of converters are well known to the person skilled in the art.
FIG. 1 shows a possible architecture of a full-bridge converter with current doubler 20.
In particular, such a converter comprises a transformer T with a primary winding T1 and a secondary winding T2. In particular, the transformer T can be modelled as a first inductor coupled in parallel to the primary winding T1, which represents the magnetizing inductance of the transformer T, a second inductor coupled in series to the primary winding T1, which represents the leakage inductance of the transformer T, and an ideal transformer with a given turn ratio N:1.
In the example considered, the converter 20 receives a DC voltage Vin at an input through two input terminals 202 and GND1, and supplies a voltage Vo at an output through two output terminals 206 and GND2.
In general, the voltage Vin may also be obtained from an AC input current, for example, via an input stage comprising a rectifier, such as a diode or a diode bridge and possibly one or more filters, such as capacitors.
In the example considered, the converter 20 comprises, on the primary side of the transformer T, an H-bridge (or full-bridge), which comprises four electronic switches Q1, Q2, Q3, and Q4, such as n-channel MOSFETs (Metal-Oxide-Semiconductor Field-Effect Transistors), which can be used for selectively connecting the two terminals of the primary winding T1 of the transformer T to the line 202 or to the ground GND1.
In particular, in the example considered, the electronic switches Q1 and Q2 are coupled in series between the lines 202 and GND1 and also the electronic switches Q3 and Q4 are coupled in series between the lines 202 and GND1. Furthermore, the intermediate point A between the switches Q1 and Q2 is coupled to the first terminal of the primary winding T1, and the intermediate point B between the switches Q3 and Q4 is coupled to the second terminal of the primary winding T1. Consequently, an H-bridge (or full-bridge) comprises two half-bridges.
The converter 20 comprises, on the secondary side of the transformer T, a current doubler, which includes two inductors Lout1/Lout2, two diodes S1/S2, and an output capacitor Cout. The person skilled in the art will appreciate that, instead of the diodes S1/S2, in general, other electronic switches may also be used, such as, for example, n-channel MOSFETs, which are driven in an appropriate way.
In particular, the above circuit has the purpose of transferring both of the half-waves of the oscillation at the secondary T2 of the transformer T to the capacitor Cout. For this purpose, a first terminal of the secondary winding T2 is coupled, through the inductor Lout1, to the positive terminal of the capacitor Cout, i.e., the terminal 206 and the second terminal are coupled through the inductor Lout2 to the positive terminal of the capacitor Cout. The negative terminal of the capacitor Cout that represents the second ground GND2, which on account of the insulating effect of the transformer T is preferably different from the ground GND1 and is consequently represented with a different ground symbol, is coupled through the diode S2 to the first terminal of the secondary winding T2 and through the diode S1 to the second terminal of the secondary winding T2.
Consequently, during a positive half-wave, the current flows through the inductor Lout1, the capacitor Cout, and the diode S1, and during a negative half-wave, the current flows through the inductor Lout2, the capacitor Cout, and the diode S2.
Finally, the voltage on the capacitor Cout corresponds to the voltage Vo that is supplied through the output terminals 206 and GND2.
Typically, the converter 20 comprises a control unit (not shown) that drives the switches Q1, Q2, Q3, and Q4 (and possibly the switches S1/S2). The possible forms of driving such a full-bridge converter are well known in the art, and a possible type of driving is described, for example, in the paper by Douglas Sterk, et al., “High Frequency ZVS Self-driven Full-Bridge Using Full Integration of Magnetics”, Applied Power Electronics Conference and Exposition, 2005 (APEC 2005), the contents of which are incorporated herein for reference for this purpose.
For example, these DC-DC converters are frequently driven in resonant or quasi-resonant mode, since this offers a high efficiency of conversion for input voltages Vin (referred to as “bus voltages”) higher than 12 V, and consequently these converters are frequently used in applications where the power bus voltages are, for example, 24 V, 48 V, or 400 V.
For example, frequently a first half-bridge is driven with a given duty cycle and the other half-bridge is switched at the same frequency and with a known delay, or time-shift, with respect to the first. These converters offer high efficiency because an appropriate series and/or parallel resonant network provided by dual elements (capacitance and inductance) is designed so that the switchings of the switches Q1-Q4 and preferably also the switches S1/S2 occur in conditions of zero voltage drop across them (zero-voltage switch—ZVS) and possibly also in condition of zero current flowing through them (zero-current switch—ZCS). For these conditions to be present, an accurate selection of the switching frequencies and of the time-shift is usually necessary.
Frequently, the specifications for these converters are stringent in terms of accuracy of the output voltage Vo. Consequently, to minimize the oscillations given by the resonance or by the switching frequency, the current doubler is provided with a second-order low-pass filter, i.e., the inductors Lout1, Lout2 and the capacitor Cout.
In general, the current-doubler rectifier (which comprises the transformer T, the inductors Lout1/Lout2, the capacitor Cout, and the switches S1/S2) can also be used in other converters. For example, the paper by Jian Sun, et al., “An Improved Current-Doubler Rectifier with Integrated Magnetics”, Applied Power Electronics Conference and Exposition, 2002 (APEC 2002), shows that, instead of an H-bridge, a half-bridge and two capacitors may also be used. The person skilled in the art will appreciate that the choice of using a full-bridge or a half-bridge typically depends upon the power supplied by the converter 20.
Consequently, the resonant circuit of the converter 20 comprises the inductances of the transformer T, the inductors Lout1/Lout2, the capacitor Cout, and the capacitances of the switches Q1-Q4 of the half-bridge or full-bridge. The documents cited previously show in this context the possibility of integrating the transformer T and the inductances Lout1, Lout2 in a single magnetic component. For example, in FIG. 2a of the paper by Jian Sun, winding of the primary is performed in the magnetic circuit (generally provided by ferromagnetic material) on the central leg where the secondary winding is present, coupling therewith and thus providing the transformer, whereas on the lateral legs of the core the resulting magnetic flux is captured by the auxiliary windings of the secondary, thus providing the inductances Lout1 and Lout2 (inductances L1 and L2 in the paper). The remaining FIGS. 2b and 2c of the paper show similar implementations.
Consequently, it is possible to obtain in a single magnetic component the transformer T and the inductances Lout1, Lout2 according to the known art.
In general, the resonance network of the primary is given by the parasitic capacitance through the terminals of the switches of the half-bridge or full-bridge and the parasitic or leakage inductance of the transformer T. However, as explained previously, the resonance frequency is fundamental for proper operation of the circuit, and the dependence upon the leakage inductance means having a dependence of a fundamental parameter upon the variation of the manufacturing parameters of the transformer T and of the board that contains the circuit. Another disadvantage of this approach lies in the fact that the greater the leakage inductance of the transformer T, the greater also the power losses of the transformer T.
Furthermore, there are resonant converters in which proper operation of the circuit and maintenance of the ZVS conditions are ensured for a given range of values of inductance at the primary T1 and, consequently, the magnetic integration as per the known art may prove difficult because the converter may require a real inductance on the primary.